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參數(shù)資料
| 型號： | L4782TA |
| 廠商： | National Semiconductor Corporation |
| 英文描述： | 3 Channel 25W Audio Power Amplifier with Mute and Standby |
| 中文描述： | 3頻道25瓦音頻功率放大器靜音和待機 |
| 文件頁數(shù)： | 19/26頁 |
| 文件大?。?/td> | 1157K |
| 代理商： | L4782TA |
Application Information
(Continued)
Mute Mode and Standby Mode sections in the
Application
Information
section for a more detailed description of these
functions.
CLICKS AND POPS
In the typical application of the LM4782 as a split-supply
audio power amplifier, the IC exhibits excellent “click” and
“pop” performance when utilizing the mute mode. In addition,
the device employs Under-Voltage Protection, which elimi-
nates unwanted power-up and power-down transients. The
basis for these functions are a stable and constant half-
supply potential. In a split-supply application, ground is the
stable half-supply potential. But in a single-supply applica-
tion, the half-supply needs to charge up at the same rate as
the supply rail, V
. This makes the task of attaining a
clickless and popless turn-on more challenging. Any uneven
charging of the amplifier inputs will result in output clicks and
pops due to the differential input topology of the LM4782.
To achieve a transient free power-up and power-down, the
voltage seen at the input terminals should be ideally the
same. Such a signal will be common-mode in nature, and
will be rejected by the LM4782. In
Figure 4
, the resistor R
serves to keep the inputs at the same potential by limiting the
voltage difference possible between the two nodes. This
should significantly reduce any type of turn-on pop, due to an
uneven charging of the amplifier inputs. This charging is
based on a specific application loading and thus, the system
designer may need to adjust these values for optimal perfor-
mance.
As shown in
Figure 4
, the resistors labeled R
BI
help bias up
the LM4782 off the half-supply node at the emitter of the
2N3904. But due to the input and output coupling capacitors
in the circuit, along with the negative feedback, there are two
different values of R
, namely 10k
and 200k
. These
resistors bring up the inputs at the same rate resulting in a
popless turn-on. Adjusting these resistors values slightly
may reduce pops resulting from power supplies that ramp
extremely quick or exhibit overshoot during system turn-on.
PROPER SELECTION OF EXTERNAL COMPONENTS
Proper selection of external components is required to meet
the design targets of an application. The choice of external
component values that will affect gain and low frequency
response are discussed below.
The gain of each amplifier is set by resistors R
f
and R
for the
non-inverting configuration shown in
Figure 1
. The gain is
found by
Equation (5)
below:
A
V
= 1 + R
f
/ R
i
(V/V)
For best noise performance, lower values of resistors are
used. A value of 1k
is commonly used for R
i
and then
setting the value of R
f
for the desired gain. For the LM4782
the gain should be set no lower than 10V/V and no higher
than 50V/V. Gain settings below 10V/V may experience
instability and using the LM4782 for gains higher than 50V/V
will see an increase in noise and THD.
The combination of R
i
with C
i
(see
Figure 1
) creates a high
pass filter. The low frequency response is determined by
these two components. The -3dB point can be found from
Equation (6)
shown below:
f
i
= 1 / (2
π
R
i
C
i
) (Hz)
If an input coupling capacitor is used to block DC from the
inputs as shown in
Figure 5
, there will be another high pass
filter created with the combination of C
IN
and R
IN
. When
(5)
(6)
using a input coupling capacitor R
is needed to set the DC
bias point on the amplifier’s input terminal. The resulting
-3dB frequency response due to the combination of C
IN
and
R
IN
can be found from
Equation (7)
shown below:
f
IN
= 1 / (2
π
R
IN
C
IN
) (Hz)
With large values of R
oscillations may be observed on the
outputs when the inputs are left floating. Decreasing the
value of R
IN
or not letting the inputs float will remove the
oscillations. If the value of R
IN
is decreased then the value of
C
IN
will need to increase in order to maintain the same -3dB
frequency response.
(7)
HIGH PERFORMANCE CONSIDERATIONS
Using low cost electrolytic capacitors in the signal path such
as C
and C
(see Figures 1 - 5) will result in very good
performance. However, electrolytic capacitors are less linear
than other premium capacitors. Higher THD+N performance
may be obtained by using high quality polypropylene capaci-
tors in the signal path. A more cost effective solution may be
the use of smaller value premium capacitors in parallel with
the larger electrolytic capacitors. This will maintain signal
quality in the upper audio band where any degradation is
most noticeable while also coupling in the signals in the
lower audio band for good bass response.
Distortion is introduced as the audio signal approaches the
lower -3dB point, determined as discussed in the section
above. By using larger values of capacitors such that the
-3dB point is well outside of the audio band will reduce this
distortion and improve THD+N performance.
Increasing the value of the large supply bypass capacitors
will improve burst power output. The larger the supply by-
pass capacitors the higher the output pulse current without
supply droop increasing the peak output power. This will also
increase the headroom of the amplifier and reduce THD.
SIGNAL-TO-NOISE RATIO
In the measurement of the signal-to-noise ratio, misinterpre-
tations of the numbers actually measured are common. One
amplifier may sound much quieter than another, but due to
improper testing techniques, they appear equal in measure-
ments. This is often the case when comparing integrated
circuit designs to discrete amplifier designs. Discrete transis-
tor amps often “run out of gain” at high frequencies and
therefore have small bandwidths to noise as indicated below.
20081199
Integrated circuits have additional open loop gain allowing
additional feedback loop gain in order to lower harmonic
distortion and improve frequency response. It is this addi-
tional bandwidth that can lead to erroneous signal-to-noise
measurements if not considered during the measurement
process. In the typical example above, the difference in
bandwidth appears small on a log scale but the factor of 10in
bandwidth, (200kHz to 2MHz) can result in a 10dB theoreti-
cal difference in the signal-to-noise ratio (white noise is
proportional to the square root of the bandwidth in a system).
L
www.national.com
19
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